Methods and apparatus for generating and utilizing training codes in a space division multiplexing communication system

ABSTRACT

A communication system is disclosed that comprises first and second communication apparatus. The first communication apparatus is provided with a transmitter for each of the transmission paths, which are capable of sending at least part of a communication signal to the second communication apparatus. The first communication apparatus moreover comprises training mechanisms for generating a training code to be sent to the second communication apparatus enabling a receiver to match a received signal to a corresponding transmitted signal. According to the invention, a training code is used with at least nearly ideal cyclic auto-correlation properties such that its cyclic auto-correction function is at least nearly zero for all cyclic shifts. The transmitter concurrently sends a training code in a mutually shifted manner, while the receiver is capable of performing a cyclic auto-correlation with respect to a received training signal.

CROSS-REFERENCE TO RELATED APPLICATION

This application claims priority of European Patent Application No.00310085.6, which was filed on Nov. 13, 2000.

BACKGROUND OF THE INVENTION

Communication systems, especially of a non-wired type, share a commonfaith in that they have to cope with the available bandwidth over thetransmission path used. Numerous techniques have been developed throughthe years to utilize the available bandwidth as efficient as possible inorder to enhance the bit rate over the transmission path. One of thesetechniques is so called space division multiplexing (SDM) by which acommunication signal is fed and divided over a number of separatetransmission paths in parallel. The communication means of the typedescribed in the opening paragraph employ this technique and for thatpurpose are equipped with transmission and reception means for everytransmission path which is used for the exchange of the communicationsignal.

A problem encountered with SDM in a non-wired environment is that thesignals sent by the different transmitters are likely to interfere witheach other such that each receiver not only receives a signal from theassociated transmitter but also from the other transmitters. In a simplecase of two transmitters and two receivers the signals r₁ and r₂received by the two receivers may be represented as follows:

r₁ = h₁ ⋅ x₁ + h₂ ⋅ x₂ r₂ = h₃ ⋅ x₁ + h₄ ⋅ x₂where x₁ and x₂ are the signals sent by both transmitters and h_(1.4)are still unknown constants representing distortion and attenuationfactors as well as other environmental and atmospheric influencesexerted on the transmitted signals during transport. More generally thismay be represented as: r_(i)=H_(i).x_(i), where H_(i) is the appropriatecolumn of a n-dimensional matrix containing the constants h_(j), saiddimension n being equal to the number of transmission paths used. Ifthese constants are known, the system will be able to satisfactorilydecode the signals received by the receivers and to derive the originalsignals x_(i) out of them. Accordingly it is necessary to teach thesystem these factors before information is being sent. To this end thesystem is provided with training means which are capable of issuingknown training codes which are packed into one or more training symbolswhich precede the information to be sent. Based on these training codes,the reception means will be able to calculate the constants h_(j) and toapply these constants to the information which follows.

A straightforward training scheme would be to use a predeterminedtraining symbol and to send at least one such symbol by each transmitterconsecutively while the other transmitter(s) are inactive. In thismanner the receiver may calculate the first column of the above matrixfrom the training symbol sent by the first transmitter, the secondcolumn from the training symbol sent by the second transmitter, and soon. Hence, a system using n transmission paths will require a minimum ofn training symbols to recover the constants h_(j). This training lengthis a serious problem for high rate wireless packet transmission links,because of the associated overhead which reduces the net data rate. Forinstance, at 100 Mbps, a 1000 byte packet has a duration of 80 μs. Usingthe same 4 μs symbol duration as the IEEE 802.11a standard, a systemusing 4 transmission paths would take a minimum training time of 16 μs,which means a significant overhead of 20%. The training overhead growswith bit rate and number of transmission paths so it partially reducesthe benefits of using SDM.

The present invention has inter alia for its object to provide acommunication system which the training time may be reduced considerablycompared to the above. It is a further object of the invention toprovide such a communication system with improved frame detection andfrequency synchronization.

SUMMARY OF THE INVENTION

The present invention relates to a communication system comprising firstcommunication means, second communication means and a first transmissionpath as well as at least one further transmission path between saidfirst and said second communication means, in which at least the firstcommunication means are provided with transmission means for each ofsaid transmission paths, which are capable of sending at least part of acommunication signal to the second communication means, in which atleast the second communication means comprise reception means for eachof said transmission paths, which are capable of receiving at least partof said communication signal, and in which at least the firstcommunication means comprise training means for generating a trainingcode to be sent to the reception means enabling the reception means tomatch a received signal to a corresponding transmitted signal.

To this end a communication system of the kind described in the aboveparagraph is according to the invention characterized in that thetraining means are capable of generating a training code with at leastnearly ideal cyclic auto-correlation properties such that its cyclicauto-correlation function is at least nearly zero for all cyclic shifts,in that the transmission means are capable of concurrently sending saidtraining code in a mutually shifted manner and in that the receptionmeans are capable of performing a cyclic auto-correlation with respectto a received training signal. Because of the cyclic auto-correlationproperties of the training code applied in accordance with the inventionit is achieved that the auto-correlation performed on the receivedtraining signal leaves no or hardly no by-products. Instead, with idealauto-correlation properties, the output of the auto-correlationoperation at the reception side will be either n.h_(i) or zero, whereh_(i) is one of the constants to be estimated and n is a knownnormalization factor. In fact the auto-correlation properties need notbe absolutely ideal but may deviate to a small extent from the idealsituation, leaving sufficient certainty to derive the requiredconstants. Because the transmitters send the training code concurrentlyin time, the invention, in theory, requires only one training symbol'sduration to recover all constants h_(i).

The auto-correlation operation requires that the training code bemultiplied by all itself and by all its cyclic shifts to render theabove product. This may be effected by sending not only the trainingcode but also its cyclic shifts to the reception means and multiplyingthese codes with the training code generated by the reception means. Ina preferred embodiment, however, the communication system according tothe invention is characterized in that the reception means are capableof generating the cyclic shifts of a received training code and tocorrelate these with said training code. In this case, only the trainingcode has to be sent and this code as well as all cyclic shifts aregenerated at the reception side. It not only limits this embodimenttransmission time and hence training time, it also avoids any distortionor other noise of the correlation products.

A specific embodiment of the communication system according to theinvention is characterized in that the training code comprises aconcatenation of the rows of a Fourier matrix. Such a concatenation isgenerally referred to as a Frank and Zadoff-Chu sequences and happen tohave ideal auto-correlation properties. This renders these sequencesextremely suitable for use in a communication system in accordance withthe present invention.

The length of the training code is preferably equal to the number ofconnection paths used or a integer multiple thereof.

In yet another specific embodiment, the communication system accordingto the invention is characterized in that the training code y is derivedfrom a maximal length sequence x with an uneven length L, having anauto-correlation of −1 for all cyclic shifts, such that at leastapproximately y=x+j/ωL. A maximal length sequence has anauto-correlation of −1 for all cyclic shifts. A sequence of this kindwith an uneven length can be modified into the above code y, which issuitable for use in the communication system according to the invention.

In order to deal with possible delay spread of the training signalsreceived by the second communication system, a further embodiment of thecommunication system according to the invention is characterized inthat, during operation, the training codes are preceded and followed bya dummy code. The dummy codes are taken sufficiently long to avoidsubstantial overlap between the auto correlation output signals duringthe training stage, so that the constants may be derived unambiguously.

In a further preferred embodiment, the communication system according tothe invention is characterized in that the training means comprise apre-correction filter for processing the training code. A pre-correctionfilter as used in this embodiment is not absolutely necessary, butwithout it the reception means generally will have to perform acorrection. The additional complexity and possible signal noise may besaved by pre-correcting the training signals before they are sent.

In a further preferred embodiment, the communication system according tothe invention is characterized in that the training means comprisestorage means for storing one or more training codes. The transmittedtraining signal can be pre-calculated and stored in memory to avoid thecomplexity of a separate pre-correction filter which would otherwise beused merely during the training phase. The pre-corrected training codesare for instance stored in a lookup table easily accessible for thetraining means. During the training stage, these codes merely need to beread out so that no complex circuitry is needed for generation orpre-correction of the codes concerned.

In still a further preferred embodiment of the communication systemaccording to the invention not merely one training code is issued duringthe training phase but instead the training means, during operation,issue a number of at least substantially identical training codes and inthat the receiving means comprise summation means to average thereceived training codes. By issuing a repetition of training codes andadding the auto-correlation output over time, a significant improvementin signal to noise ratio may be obtained.

The use of a repetitive pattern of the same code as training signal,moreover provides an excellent frame detection trigger. To this end, afurther specific embodiment of the communication system according to theinvention is characterized in that the training means, at least duringoperation, issue at least substantial training codes at a substantiallyfixed interval and in that the reception means are provided with autocorrelation means for correlating a received signal with one or moresignals received after a delay corresponding to said interval or aninteger multiple thereof. Because of the ideal auto-correlationproperties of the code, the correlation outputs consist only of the sumof n₁ auto-correlation points, whereas all cross-correlation productsbetween signals coming from different transmitters are zero. If there isa valid training signal at the input of the reception means, then thecorrelator output values are proportional to the total received power oneach reception means. The phase of the correlator outputs is equal to2Bf_(o)T_(c), where f₀ is the frequency offset between transmissionmeans and the reception means and T_(c) the time interval betweensuccessive training codes. After summing all correlator outputs, a startof a frame is detected by looking if the magnitude of the output signalexceeds some threshold. The value of the threshold is a tradeoff betweenthe probability of a false alarm and the detection probability. If astart-of-transmission is detected, then the phase of the summedcorrelator outputs is a measure for the carrier frequency offset thatcan be fed to a frequency correction circuit. The frequency offset f_(o)is given by the output phase divided by 2BT_(c).

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will be described in further detail below along the linesof a specific example and with reference to the accompanying drawings.In the drawings:

FIG. 1 is a schematic representation of a specific embodiment of acommunication system in accordance with the invention;

FIG. 2 is the in phase part of the correlation output of a receiver uponreception of a training signal within the system of FIG. 1 for twotransmit antennas;

FIG. 3 is the windowed correlation output of FIG. 2;

FIG. 4 is the in phase channel coefficient for different sub-carriers,both estimated and actual, in the system of FIG. 1;

FIG. 5 is a first embodiment of a frame detection and frequencysynchronization circuit for use in a communication system according tothe invention;

FIG. 6 is a second embodiment of a frame detection and frequencysynchronization circuit for use in a communication system according tothe invention; and

FIG. 7 is a third embodiment of a frame detection and frequencysynchronization circuit for use in a communication system according tothe invention.

Like parts have been given the same reference numerals throughout thefigures.

DETAILED DESCRIPTION

The communication system depicted in FIG. 1 comprises firstcommunication means 10 with transmission means which are capable ofsending a communication signal in a wireless communication network aswell as second communication means 20 which comprise reception meanswhich are capable of receiving said communication signal. In thisexample, the communication means are primarily intended for dataexchange and operate based on a TCP/IP or any other suitable packageddata transmission protocol.

Especially with data transmission the finite available bandwidthpresents an everlasting limitation. In order to cope with thislimitation, several multiplexing techniques have been developed,including space division multiplexing (SDM) as used in this example.According to the technique, the transmission means comprise multipletransmitters 11 . . . 44 which are concurrently used for thetransmission of the communication signal over multiple transmissioncarriers, whereas also multiple reception means 31 . . . 34 areavailable for receiving the signals sent. The different transmissionmeans 11 . . . 14 and reception means 31 . . . 34 each comprise theirown antenna, which is depicted schematically in the drawing. Moreover, adivision of the signal over several orthogonal frequency bands iseffected in order to further enhance the capacity of the system,generally known as orthogonal frequency division multiplexing (OFDM).The communication system is distributed over the different sub-carriersand frequencies to maximize the data throughput capacity of theconnection.

SDM used in wireless data transmission presents a complication in thatthe signals of the different transmission means 11 . . . 14 willinevitably be received by all reception means 31 . . . 34. For fourtransmitters 11 . . . 14, like in this example, sending signals x₁ . . .x₄ respectively on a specific sub-carrier, the signals r₁ . . . r₄received by the different receivers 31 . . . 34 are given by:

r₁ = h₁ ⋅ x₁ + h₂ ⋅ x₂ + h₃ ⋅ x₃ + h₄ ⋅ x₄r₂ = h₅ ⋅ x₁ + h₆ ⋅ x₂ + h₇ ⋅ x₃ + h₈ ⋅ x₄r₃ = h₉ ⋅ x₁ + h₁₀ ⋅ x₂ + h₁₁ ⋅ x₃ + h₁₂ ⋅ x₄r₄ = h₁₃ ⋅ x₁ + h₁₄ ⋅ x₂ + h₁₅ ⋅ x₃ + h₁₆ ⋅ x₄

in which h_(1 . . . 16) represent transmission factors or channelcoefficients which take into account the spatial displacement of thedifferent transmitters and receivers as well as different environmentalinfluences giving rise to, for instance, attenuation and distortion ofthe signal. Because these factors are not known and may change from timeto time, they need to be recovered for each data package being sent.This process is generally referred to as channel training and consistsof the transmission of a number of known data signals which enable thereception means to recover the transmission factors h_(1 . . . 16). Tothis end, the transmission means are equipped with training means 21 . .. 24. Although these training means are indicated separately for alltransmitters, they may be shared among the transmitters in order to savecircuitry.

The training means may comprise a processor unit capable of calculatingthe appropriate training signals, but in this example, simply consist ofa look-up table which has been filled in advance with suitable trainingcodes. In accordance with the present invention, these training codesare chosen to have at least nearly ideal auto-correlation properties,meaning that their auto-correlation product is zero for all cyclicshifts and non-zero for itself. In this example, the so-called Frank andZadoff-Chu sequence is taken as training code. A Frank code is obtainedby concatenating all rows or columns of a discrete Fourier matrix. The 4by 4 discrete Fourier matrix F, for example, is given by:

$F = \begin{bmatrix}1 & 1 & 1 & 1 \\1 & j & {- 1} & {- j} \\1 & {- 1} & 1 & {- 1} \\1 & {- j} & {- 1} & j\end{bmatrix}$

By concatenating the rows of this matrix, the following sequence c isobtained having perfect auto-correlation properties:

c={1, 1, 1, 1, 1, j, −1, −j, 1, −1, 1, −1, j, −1, j}

The auto-correlation function of this code renders zero for all itscyclic shifts and equals 16 for the non-shifted auto-correlation. Inthis example, use is made of a code c with a length (16) equal to aninteger multiple (4) of the number of sub-carriers (4).

The same training code c is supplied to all transmitters 11 . . . 14,but in a cyclically shifted fashion. This means that, for instance, thefirst transmitter 11 transmits the original code c, the secondtransmitter 12 transmits the code cyclically shifted over one digit, thethird transmitter 13 transmits the code c shifted over two digits, andso on. In order to compensate for any distortion during transmission, apre-correction filter is applied to the code c so that its spectrum willbe the same as that of the OFDM signal. It is not absolutely necessaryto add a pre-correction but, without it, the receivers need to make acorrection which would make the channel estimates more noiser forsub-carriers corresponding to low code spectral values of thetransmitted code. The corrected training signal is precalculated andstored in the look-up table of the training means 21 . . . 24 to avoidthe complexity of a separate pre-correction filter which would otherwiseonly be used for the training phase.

The second communication means 20 comprise separate receivers 31 . . .34 with associated antennas for all transmitters 11 . . . 14 of thefirst communication means 10. These receivers 31 . . . 34 will eachreceive the training codes c coming from all transmitters 11 . . . 14.The receivers are each coupled to correlation means 41 . . . 44 whichperform a cyclic correlation of the code c with part of the signalreceived by the associated receiver 31 . . . 34 of a length equal to thecode c. If the training length is more than twice the code length, thenthe receivers 31 . . . 34 first sum or average an integer number ofparts of the training signal with a length equal to that of the code c.The averaged signal is then used to perform the cyclic correction.Because of the ideal cyclic auto-correlation properties of the trainingcode c, as used in the communication system according to the invention,at all times, only the signal of one of the transmitters 11 . . . 14,matching the code at that time applied by the cyclic correlation means41 . . . 44, will deliver a non-zero output whereas the signals receivedfrom the other transmitters will be cancelled. By cyclically shiftingthe code c, all coefficients may be recovered in this manner, one afterthe other.

FIG. 2 shows an example of the cyclic correlation output for twotransmitters. The cyclic correlation output shows separate impulseresponses for different transmission antennas, which are separated by adelay equal to the cyclic shift applied in the subject transmitter. Thecode in this example is a length 64 Frank sequence that is two timesover sampled, so there are 128 samples in one code length. All receiverswill see a different cyclic correlation function from which they canextract the channel information.

To estimate a specific channel coefficient h_(j) the reception meanscomprise a filter 51 . . . 54 which isolates the impulse response of thesubject transmitter by multiplying the cyclic correlation output of thecorrelator 41 . . . 44 by a window function which is non-zero only atthe desired impulse response. The window has smooth roll-off regions inorder to minimize errors in the estimated channel frequency response.The roll-off factor is a compromise between frequency leakage and delayspread robustness; a larger roll-off region gives less frequency leakagebut it also attenuates more impulse response components whose pathdelays fall within the roll-off region. FIG. 3 shows an example of awindowed correlation function based on the correlator output shown inFIG. 2.

The windowed impulse response, taken at the output of the filter 51 . .. 54, is cyclically shifted back by a second correlator 61 . . . 64 tocompensate for the cyclic shift applied to the transmitter. Finally, thechannel frequency response is found by calculating a Fast FourierTransform (FFT) over the windowed impulse response, taken at the outputof the second correlator 61 . . . 64. To this end, the reception meanscomprise a FFT-filter 71 . . . 74. The FFT-filter 71 . . . 74 outputsresemble the channel coefficient values of all OFDM sub-carriers for oneparticular transmitter-receiver pair. The estimated channel frequencyresponse for all sub-carrier is drawn as curve a in FIG. 4, whereascurve b represents the actual values. At the lower sub-carriers, thereis minor difference with the actual channel frequency response, which isassumed to be introduced by the windowing operation of the first filter51 . . . 54.

Although the system of this example uses OFDM together with SDM, theinvention may as well be used for SDM on its own, i.e. with merely asingle carrier. If peak power is limited, which is often the case, thensingle carrier SDM training might require a separate training signal pertransmit antenna instead of a common signal source. The training signalhas to be long enough to get sufficient signal-to-noise ratio (SNR) atthe receiver. Instead of sending separate training signals over theantennas one by one, one long the training signal may then betransmitted simultaneously on all antennas. In the receiver, the sameprocedure outlined above can be followed up to the cyclic correlation.The outputs of the cyclic correlation directly resemble the desiredchannel coefficients that are needed for a single carrier SDM receiver.

Theoretically, the training time need not be longer than the duration ofthe training code c. In practice however, a repetition of the trainingcode is used in order to realize sufficient SNR at the receiver's end.Moreover, in order to deal with inevitable delay spread, the code c ispreferably extended with a small dummy code preceding and following it.The addition of said dummy codes ascertains that the impulse responsesassociated with the different transmitters do not overlap mutually. Thisextended code is then stored in the look-up table of the training means21 . . . 24. Even with this extension, the total training time need notbe much longer than the duration of a training signal with sufficientSNR, whereas the prior art systems all require a multiple thereof.

The fact that, in a communication system according to the invention, thetraining signal consists of a repetitive pattern with at least nearlyideal auto-correlation properties may also advantageously be used forframe detection and the detection of the carrier frequency offset. FIG.5 shows a block diagram of a system that detects the start of atransmission and that estimates the carrier frequency offset in acommunication system like that of FIG. 1 but with two transmitters andreceivers. For each receiver 31, 32, the reception means comprise delaymeans 81, 82 to add a delay time T_(c) to the received signal, whereT_(c) is equal to the duration of the training code c, and conversionmeans 91, 92 to convert the delayed signal to a conjugated replica ofit. This T_(c) seconds delayed and conjugated replica is fed tocorrelation means 101, 102. Because of the ideal auto-correlationproperties of the code c, the correlation outputs consist only of thesum of n_(t) auto-correlation points, whereas all cross-correlationproducts between signals coming from other transmitters are zero.

If there is a valid training signal at the input of the receivers, thenthe correlator output values are proportional to the total receivedpower on each receiver. The phase of the correlator outputs is equal to2B.f_(o).T_(c), where f_(o) is the frequency offset between transmitterand receiver. If all transmitters and receivers from one SDM terminalshare the same reference oscillator, only a single frequency offset hasto be estimated and corrected. The correlator outputs are added at 105and fed to summation means 110 to average the total in time. A start ofa frame is detected by means of a comparator 120 which detects whetherthe magnitude of the absolute value 115 of the output signal of thesummation means 110 exceeds some threshold *. The value of thethreshold * is a tradeoff between the probability of a false alarm andthe detection probability. If a start-of-transmission is detected, atrigger is raised at the output 130 of the circuit and at the same timethe actual phase v of the summed correlator outputs is a measure for thecarrier frequency offset. The frequency offset f_(o) is given by theoutput phase v divided by 2B.f_(o).T_(c) and may be derived at theoutput 140 of calculation means 125 in order to be fed to a frequencycorrection circuit.

FIG. 6 shows a block diagram of an SDM synchronization system that findsthe symbol timing based on knowledge of the transmitted training codesc. Each of the received signals is passed through a matched filter 151,152 with an impulse response equal to the conjugated and time-reversedcode c. Hence, the matched filter output shows the correlation betweenthe received signal and c. As there is no knowledge of the carrier phaseat this stage, the power of the correlation outputs is calculated bytaking the square of the output value with suitable means 160. The poweroutputs of all receivers are added at 165 to improve the signal-to-noiseratio (SNR) of the detector output. A further SNR enhancement isobtained by adding signal components which are spaced apart by T_(d)seconds 170, where T_(d) is equal to the cyclic time shift applied tothe codes c in the SDM transmitters. The eventual output is fed tocomparator 175 which selects the largest peak. The occurrence of thispeak is a trigger that can be used for symbol timing.

Besides adding and averaging the training signals of multipletransmitters and receivers, moreover averaging over several trainingcodes is possible to further enhance the SNR. To this end, an additionalFinite Impulse Response (FIR) filter 172 is added to the circuit of FIG.6 that adds samples which are spaced apart by multiple code lengths ofT_(c) seconds, see FIG. 7. As the training signal consists of arepetition of the same code c, this further improves the SNR. Thecircuit of FIG. 7, moreover, comprises an extra FIR filter 174 that sumsover multiple multipath components with a spacing of one sample intervalT_(s). The number of taps L of this last FIR filter should preferably besuch that L. T_(s) is equal to the maximum tolerable delay spread thatthe receiver can tolerate. This structure gives an enhanced SNRperformance in the case of multipath fading channels. In addition, itimproves the maximum tolerable delay spread.

Although the invention has been described in further detail withreference to merely a number of embodiments, it will be appreciated thatthe invention is by no means limited to the examples given. On thecontrary a skilled person will be able to arrive at numerous differentembodiments and variation without departing from the scope and spirit ofthe present invention. As such he may avail himself of other codes withideal or nearly ideal cyclic auto-correlation properties to be used astraining codes. Besides the aforementioned Frank and Zadoff-Chusequences, for instance maximum length sequences, having anauto-correlation of −1 for all cyclic shifts, may be used as a basis ofsuch an alternative code.

1. A communication system having first communication means, secondcommunication means and a first transmission path as well as at leastone further transmission path between said first and said secondcommunication means, in which at least the first communication means areprovided with transmission means for each of said transmission paths,which are capable of sending at least part of a communication signal tothe second communication means, in which at least the secondcommunication means comprise reception means for each of saidtransmission paths, which are capable of receiving at least part of saidcommunication signal, wherein the first communication means comprises: atraining generator that generates a training code to be sent to thereception means enabling the reception means to match a received signalto a corresponding transmitted signal, wherein the training generator iscapable of generating a training code with at least nearly ideal cyclicauto-correlation properties such that its cyclic auto-correlationfunction is at least nearly zero for all cyclic shifts, in that thetransmission means are capable of concurrently sending said trainingcode in a mutually shifted manner and in that the reception means arecapable of performing a cyclic auto-correlation with respect to areceived training signal.
 2. Communication system according to claim 1,wherein the reception means are capable of generating the cyclic shiftsof a received training code and to correlate these with said trainingcode.
 3. Communication system according to claim 1, wherein the trainingcode comprises a concatenation of the rows of a Fourier matrix. 4.Communication system according to claim 3, wherein the training code hasa length which is equal to the number of transmission paths or aninteger multiple thereof.
 5. Communication system according to claim 1,wherein the training codes are preceded and followed by a dummy codeduring operation.
 6. Communication system according to claim 1, whereinthe training generator comprises a pre-correction filter for processingthe training codes.
 7. Communication system according to claim 1,wherein the training generator comprises storage means for storage oneor more training codes.
 8. Communication system according to claim 1,wherein the training generator during operation, issue a number of atleast substantially identical training codes and in that the receivingmeans comprise summation means to average the received training codes.9. Communication system according to claim 1, wherein the traininggenerator at least during operation, issue at least substantial trainingcodes at a substantially fixed interval and in that the reception meansare provided with auto-correlation means for correlating a receivedsignal with one or more signals received after a delay corresponding tosaid interval or an integer multiple thereof.